1. Field of the Invention
This invention pertains to method and apparatus for determining and/or compensating for a time drift between sample clocks of a transmitter and a receiver in conjunction with transmission of plural modulated signal carriers over an air or radio interface.
2. Related Art and other Considerations
Various methods for the transmission of digital signals, such as digital video broadcasting (DVB) and digital audio broadcasting (DAB) signals, are known. One method typically used for such transmissions is the orthogonal frequency division multiplexing (OFDM) method wherein a plurality of modulated signal carriers are used broadcast the signals. Multicarrier modulation schemes as the OFDM are typically used in systems wherein the time dispersion thereof is much greater than the employed bit duration. In multicarrier modulation schemes, the modulated signal carriers are sampled before being transposed in the frequency domain by means of a fast fourier transformation (FFT) for signal separation.
Orthogonal Frequency Division Multiplexing (OFDM) will also be used in wireless local area networks (WLAN) in the 5 GHz band such as specified in Europe, the U.S. and Japan. The European WLAN standard is HIgh PErformance Radio Local Area Network type 2 (HIPERLAN/2), which is being developed by the ETSI Project BRAN (broadband radio access network). The North American and the Japanese standards are expected to have very similar physical layers.
The physical layer of HIPERLAN/2 is described in ETSI TS 101 475 V1.2.2 (2001–02), Broadband Radio Access Networks (BRAN); HIPERLAN Type 2; Physical (PHY) layer, ETSI document. The air interface of HIPERLAN/2 is based on time-division duplex and dynamic time-division multiple access. All data units in every transport channel which are transmitted via the physical layer of HIPERLAN/2 are bursts.
The frame structure 18 and an exemplary burst 20 is depicted in FIG. 1. The frame includes broadcast control channel, a frame control channel, dowrilink traffic, uplink traffic, and random access channels. As shown in FIG. 1, each burst 20 comprises a preamble 22 and one or several Protocol Data Units PDUs 241, 242, 243, . . . 24n. Further details of preamble 22 and the PDUs 241, 242 . . . are illustrated in FIG. 2. Preamble 22 includes, e.g., a cyclic prefix 221 and two training symbols 222, 223. Each PDU 24 is a data unit of several bytes tat comes from the medium access control (MAC) layer. The physical (PHY) layer converts the PDU to one or more OFDM symbols, the number of symbols depending on the link adaptation mode. While FIG. 1 shows an example PDU 24 as comprising plural OFDM symbols, for sake of simplicity FIG. 2 shows its PDUs as comprising a cyclic prefix 25 and one data symbol 26. For example, in FIG. 2 PDU 241 has a cyclic prefix 251 and data symbol 261.
OFDM modulation is performed with a 64 point IFFT (Inverse Fast Fourier Transform) at a sample rate of 20 MHz, which gives a subcarrier spacing of 312.5 kHz and a symbol duration of 3.2 μs. The cyclic prefix 25 put in front of each symbol 26 is 800 ns, giving a total symbol length of 4 μs. Of the 64 subcarriers, only 52 subcarriers are used, of which 48 carry data and 4 are pilots.
The physical layer provides several link adaptation modes to accommodate for various channel conditions. Each mode comprises a combination of a subcarrier modulation scheme and a forward error correction code rate. The primary modulation schemes are BPSK, QPSK, 16QAM and 64QAM. The primary code rates are ½, 9/16, ¾.
In order to perform coherent demodulation, a receiver must synchronize with the transmitter both in time and frequency. Due to frequency differences between transmitters and receivers in such systems, the demodulated signal carriers can exhibit frequency offsets. An estimate of the channel must also be made.
The preambles 22 in the data stream facilitate, e.g., an assessment of frequency offsets of received signals and frequency synchronization. In case of the OFDM, the two identical OFDM symbols 222, 223, also referred to as C64, are inserted between cyclic prefix 21 (C32) and the actual data stream, e.g., the PDUs 24. This so-called C- preamble shown in FIG. 2 is used, e.g., for a channel estimation in the demodulation process of the multicarrier signals. Thus, an estimate of the channel is performed at the beginning of the burst with the help of the preamble, which contains the training symbols.
In HIPERLAN/2 the 20 MHz sample clock in both the transmitter and the receiver is free-running with a relative accuracy of +/−20 parts per million (ppm). The worst case scenario is therefore a +/−40 ppm offset between the sample clock frequencies in the transmitter as compared to the receiver. This potential +/−40 ppm offset causes a timing error between the transmitter and the receiver which increases with time, i.e. a timing drift. This timing drift can cause a significant disturbance in the coherent demodulation at the end of a long burst, even if the receiver is synchronized at the beginning of the burst.
The timing drift causes some undesired effects, namely: (1) phase rotation in the frequency domain; (2) inter-symbol interference; and (3) some loss of orthogonality between subcarriers. The two latter effects are not considered to influence demodulation. However, as illustrated below, the phase rotation effect can be significant.
The Fourier transform relationship of Equation 1 indicates that a time displacement Toff of a function f(t) in the time-domain gives a linear phase factor ωToff of the corresponding Fourier transform, compared to the Fourier transform F(ω) of the original function f(t).f(t−Toff)→exp(−jω·Toff)·F(ω)  Equation 1
Considering a burst with the length of a MAC frame of 2 ms, which is the longest possible burst in the HIPERLAN/2 system, the time displacement at the end of the burst will be 2 ms·40 ppm=80 ns. The phase factor for the highest used subcarrier which is at 8.125 MHz will then be ωToff=2·π·8.125 MHz·80 ns≈4.1 radians or 234 degrees. Such a phase error will, of course, render coherent demodulation impossible. A performance degradation will occur much earlier than this depending on the link adaptation mode.
There are various prior art methods for compensating for timing drift. A first such time drift compensation method is fine tuning a reference oscillator in the receiver (using an estimate of the timing drift) to render the timing drift negligible. A problem with this first method lies in the time it takes for the RF synthesizer to adjust to a new value. Typically the RF synthesizer is implemented with a phase locked loop (PLL), which needs settling times which are large compared to the typical duration of the preamble. Furthermore, a voltage-controlled reference will typically have more phase noise (fluctuations in frequency around a mean) than a free running oscillator reference.
As mentioned above, in order to do coherent demodulation the receiver has to calculate a channel estimate, initially calculated on the preamble. In accordance with a second prior art time drift compensation method, the initial channel estimate is monitored and corrected with the help of pilots or decision-directed methods during reception. In this way, there can be compensation for slow changes like timing drift. However, this approach requires complex tracking mechanism, when tracking each subcarrier separately, and the susceptibility to fading dips, when only a small number of pilots are used (HIPERLAN/2 has only 4 pilots out of 52 subcarriers).
What is needed, therefore, and an object of the present invention, is a timing drift compensation technique which is simple yet highly robust.